1. Field of the Invention
The invention relates generally to static-random-access-memory (SRAM) devices and, more particularly, to SRAM""s utilizing a four transistor design.
2. Description of the Background
To meet customer demand for smaller and more power efficient integrated circuits (ICs), manufacturers are designing newer ICs that operate with lower supply voltages and that include smaller internal subcircuits such as memory cells. Many ICs, such as memory circuits or other circuits such as microprocessors that include onboard memory, include arrays of SRAM cells for data storage. SRAM cells are popular because they operate at a higher speed than dynamic-random-access-memory (DRAM) cells, which must be periodically refreshed.
FIG. 1 is a circuit diagram of a conventional 6-transistor (6-T) SRAM cell 10, which can operate at a relatively low supply voltage, for example 1.5V-3.3V, but which is relatively large. A pair of NMOS access transistors 12 and 14 allow complementary bit values D and {overscore (D)} on digit lines 16 and 18, respectively, to be read from and to be written to a storage circuit 20 of the cell 10. The storage circuit 20 includes NMOS pull-down transistors 22 and 26, which are coupled in a positive-feedboack configuration with PMOS pull-up transistors 24 and 28, respectively. Nodes A and B are the complementary inputs/outputs of the storage circuit 20, and the respective complementary logic values at these nodes represent the state of the cell 10. For example, when the node A is at logic 1 and the node B is at logic 0, then the cell 10 is storing a logic 1. Conversely, when the node A is at logic 0 and the node B is at logic 1, then the cell 10 is storing a logic 0. Thus, the cell 10 is bistable, i.e., the cell 10 can have one of two stable states, logic 1 or logic 0.
In operation during a read of the cell 10, a word-line WL, which is coupled to the gates of the transistors 12 and 14, is driven to a voltage approximately equal to Vcc to activate the transistors 12 and 14. For example purposes, assume that Vcc=logic 1=5 V and Vss=logic 0=0 V, and that at the beginning of the read, the cell 10 is storing a logic 0 such that the voltage level at the node A is 0 V and the voltage level at the node B is 5 V. Also, assume that before the read cycle, the digit lines 16 and 18 are equilibrated to approximately Vcc-Vt. Therefore, the NMOS transistor 12 couples the node A to the digit line 16, and the NMOS transistor 14 couples the node B to the digit line 18. For example, assume that the threshold voltages of the transistors 12 and 14 are both 1 V, then the transistor 14 couples a maximum of 4 V from the digit line 18 to the node B. The transistor 12, however, couples the digit line 16 to the node A, which pulls down the voltage on the digit line 16 enough (for example, 100-500 millivolts) to cause a sense amp (not shown) coupled to the lines 16 and 18 to read the cell 10 as storing a logic 0.
In operation during a write, for example, of a logic 1 to the cell 10, and making the same assumptions as discussed above for the read, the transistors 12 and 14 are activated as discussed above, and logic 1 is driven onto the digit line 16 and a logic 0 is driven onto the digit line 18. Thus, the transistor 12 couples 4 V (the 5 V on the digit line 16 minus the 1 V threshold of the transistor 12) to the node A, and the transistor 14 couples 0 V from the digit line 18 to the node B. The low voltage on the node B turns off the NMOS transistor 26, and turns on the PMOS transistor 28. Thus, the inactive NMOS transistor 26 allows the PMOS transistor 28 to pull the node A up to 5 V. This high voltage on the node A turns on the NMOS transistor 22 and turns off the PMOS transistor 24, thus allowing the NMOS transistor 22 to reinforce the logic 0 on the node B. Likewise, if the voltage written to the node B is 4 V and that written to the node A is 0 V, the positive-feedback configuration ensures that the cell 10 will store a logic 0.
Because the PMOS transistors 24 and 28 have low on resistances (typically on the order of a few kilohms), they can pull the respective nodes A and B virtually all the way up to Vcc often in less than 10 nanoseconds (ns), and thus render the cell 10 relatively stable and allow the cell 10 to operate at a low supply voltage as discussed above. But unfortunately, the transistors 26 and 28 cause the cell 10 to be approximately 30%-40% larger than a 4-transistor (4-T) SRAM cell, which is discussed next.
FIG. 2 is a circuit diagram of a conventional 4-T SRAM cell 30, where elements common to FIGS. 1 and 2 are referenced with like numerals. A major difference between the 6-T cell 10 and the 4-T cell 30 is that the PMOS pull-up transistors 24 and 28 of the 6-T cell 10 are replaced with conventional passive loads 32 and 34, respectively. For example, the loads 32 and 34 are often polysilicon resistors. Otherwise the topologies of the 6-T cell 10 and the 4-T cell 30 are the same. Furthermore, the 4-T cell 30 operates similarly to the 6-T cell 10. Because the loads 32 and 34 are usually built in another level above the access transistors 12 and 14 and the NMOS pull-down transistors 22 and 26, the 4-T cell 30 usually occupies much less area than the 6-T cell 10.
Additional, complex steps are required to form the load elements 32 and 34 such that 4-T cells present the usual complexity versus cost tradeoff. The high resistance values of the loads 32 and 34 can substantially lower the stability margin of the cell 30 as compared with the cell 10. Thus, under certain conditions, the cell 30 can inadvertently become monostable or read unstable instead of bistable. Also, the cell 30 consumes more power than the cell 20 because there is always current flowing from Vcc to Vss through either the load 32 and the NMOS transistor 26 or the load 34 and the NMOS transistor 22. In contrast, current flow from Vcc to Vss in the cell 20 is always blocked by one of the NMOS/PMOS transistor pairs 22/24 and 26/28. Efforts to eliminate load elements 32 and 34 have lead to the development of a load-less four transistor SRAM cell as shown in FIG. 3.
FIG. 3 is a circuit diagram of a conventional load-less 4-T SRAM cell, where elements common to FIGS. 2 and 3 are referenced with like numerals. The difference between the load-less cell 36 and the cell 30 is the elimination of load elements 32 and 34 and the replacement of NMOS transistors 12 and 14 with PMOS transistors 38 and 40, respectively.
With the load-less 4-T SRAM cell of FIG. 3, like all SRAM cells, leakage currents and/or subthreshold currents are generated by transistors 22 and 26 when in the off state, and one will always be in the off state. To prevent the cell 36 from spontaneously changing state, the transistors 38 and 40 must source sufficient load current from the digit lines 16 and 18, respectively, to offset the leakage and subthreshold currents. The needed load current can vary over many orders of magnitude due to temperature and process variations. However, the load current cannot be too large because the cumulative (along the digit lines 16 and 18) load current needs to be significantly less than the cell current for proper noise margin for proper operation of the sense amps.
Wide temperature variations resulting from cold-data retention testing and burn-in testing are also causes of wide variations in leakage and subthreshold currents, thereby causing wide variations in the load current that must be sourced by transistors 38 and 40. Such testing, coupled with normal process variations, sense amp margin requirements, as well as yield requirements (e.g., read/write stability requirements, power consumption requirements, etc.) have made the manufacturing of load-less 4-T SRAM""s a difficult matter.
The present invention is directed generally to a bias generator used in conjunction with one of the word line or digit line to set the desired level of load current as a function of temperature (or test being performed) to satisfy the simultaneous constraints of yield, sense amp margin, and load current even during cold-data retention testing or burn-in.
The present invention is also directed to a method of modifying the level of current conducted by the access transistors of a load-less, four transistor memory cell when the access transistors are in an off state. The method is comprised of the step of generating a temperature dependent bias voltage and connecting that bias voltage to the gate terminals of the access transistors.
The present invention is also directed to a current-mirror-based bias generator for a load-less four transistor SRAM as well as associated methods of controlling or modifying the current conducted by the access transistors of such an SRAM. The present invention may be thought of as an adjustable temperature coefficient, bias generator that references, via a current mirror, a reference bank of SRAM cells. The bank of reference cells provides an indication of the necessary conduction characteristics (e.g., gate to source voltage) of the access transistors under various conditions. By applying a bias voltage to the word line the desired current is sourced from the digit line. The bank of reference SRAM cells inherently compensates for process variations. The adjustable temperature coefficient bias generator allows the current sourced by the digit lines to vary greatly as a result of temperature variations. Thus, the present invention compensates for both process variations and temperature variations. Those benefits, and others, will become apparent from the description of the preferred embodiment hereinbelow.